Digchip : Database on electronics components
Electronics components database

A 75 W TV Power Supply Operating in Quasi-square Wave Resonant Mode Using NCP1207 Controller

      Search in application notes:      

Application note title: A 75 W TV Power Supply Operating in Quasi-square Wave Resonant Mode Using NCP1207 Controller

Category: AC-DC Controllers
Manufacture: ON Semiconductor
Datasheet: Download this application note

AND8145/D A 75 W TV Power Supply Operating in Quasi-square Wave Resonant Mode using the NCP1207 Controller
http://onsemi.com Prepared by: Nicolas Cyr ON Semiconductor
Introduction Quasi-square wave resonant converters, also known as quasi-resonant (QR) converters, allow designing flyback Switch-Mode Power Supplies (SMPS) with reduced Electro-Magnetic Interference (EMI) signature and improved efficiency. Due to the low level of generated noise, QR SMPS are therefore very well suited to applications dealing with RF signals, such as TVs. ON Semiconductor NCP1207 is a QR controller that will ease your design of an EMI-friendly TV power supply with only a few additional components, and able to lower its standby power down to 1.0 W.
What is Quasi-Resonance?
The term quasi-resonance is normally related to the association of a real hard-switching converter and a resonant tank. While the operation in terms of control is similar to that of a standard PWM controller, an additional network is added to shape the variables around the MOSFET: current or voltage. Depending on the operating mode, it becomes possible to either switch at zero current (ZCS) or zero voltage (ZVS). Compared to a conventional PWM converter, a QR operation offers less switching losses but the RMS current circulating through the MOSFET increases and forces higher conduction losses; with a careful design, efficiency can be improved. However, one of the main advantages in favor of the quasi-resonance is the reduced spectrum content either conducted or radiated. True ZVS quasi-resonance means that the voltage present on the switch looks like a sinusoidal arch. Figure 1 shows how such a signal could look like.
Figure 1. A Truly Resonating VDS Signal on a Quasi-resonant Flyback Converter
he main problem with this technique lies in the very high voltage generated at the switch opening. Most of the time, these resonant offline designs require around 1.0 kV BVdss MOSFETs whose price is clearly incompatible with high volume markets. As a result, designers orientate their choice toward another compromise called quasi-square wave resonant power supply.
Quasi-Square Wave Resonant Converters
As we saw, true resonant operation put a constraint on MOSFET selection by imposing a high voltage at the switch opening. If we closely look at the standard hard-switching waveform (Figure 2), we can see that at a given time the drain voltage goes to a minimum. This occurs just after the core reset.
Semiconductor Components Industries, LLC, 2004
Publication Order Number: AND8145/D
F 2 @ p @ LLEAK @ CTOT
Figure 2. Hard-switching Waveforms in Discontinuous Conduction Mode (DCM)
rom Figure 2, it is possible to imagine a controller that turns a MOSFET ON until its current grows-up to the setpoint. Then it turns the MOSFET OFF until the core reset is detected (usually via an auxiliary winding). As a result, the controller does not include any stand alone clock but only detects the presence of events conditioned by load/line conditions: this is a so-called free-running operation. Converters based on this technique are often designated as Self-Oscillating Power Supplies (SOPS), valley switching converters, etc. Oscillations origins can be seen from Figure 3 arrangement where L-C networks appear.
Depending on the event, two different configurations are seen: At the switch closing, the primary current flows through the primary inductance LP but also the leakage inductance, LLEAK. When the turn-on time expires, the energy stored in LP is transferred to the secondary side of the transformer via the coupling flux. However, the leakage inductance, which models the coupling between both transformer sides, reverses its voltage and imposes a quickly rising drain voltage. The slope of this current is
where CTOT gathers all capacitors
surrounding the drain node: MOSFET capacitors, primary transformer parasitic capacitors but also those reflected from the secondary side, etc. As a result, LLEAK together with CTOT form a resonating network of natural frequency
1 2 @ p @ LLEAK @ CTOT (eq. 2) .
maximum drain voltage can then be computed using the characteristic impedance of this LC network:
VDS max + VIN ) 1 @ (VOUT ) VF) ) IP @ N LLEAK CTOT
Figure 3. A Typical Flyback Arrangement Shows Two Different Resonating Networks